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FEATURES
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LTC1538-AUX/LTC1539 Dual High Efficiency, Low Noise, Synchronous Step-Down Switching Regulators DESCRIPTION
The LTC (R)1538-AUX/LTC1539 are dual, synchronous stepdown switching regulator controllers which drive external N-channel power MOSFETs in a phase-lockable fixed frequency architecture. The Adaptive PowerTM output stage selectively drives two N-channel MOSFETs at frequencies up to 400kHz while reducing switching losses to maintain high efficiencies at low output currents. An auxiliary 0.5A linear regulator using an external PNP pass device provides a low noise, low dropout voltage source. A secondary winding feedback control pin (SFB1) guarantees regulation regardless of load on the main output by forcing continuous operation. A 5V/20mA regulator, internal 1.19V reference and an uncommitted comparator remain active when both controllers are shut down. A power-on reset timer (POR) is included which generates a signal delayed by 65536/fCLK (typ 300ms) after the controller's output is within 5% of the regulated first voltage. Internal resistive dividers provide pin selectable output voltages with remote sense capability on one of the two outputs. The operating current levels are user-programmable via external current sense resistors. Wide input supply range allows operation from 3.5V to 30V (36V maximum).
, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power is a trademark of Linear Technology Corporation.
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Maintains Constant Frequency at Low Output Currents Dual N-Channel MOSFET Synchronous Drive Programmable Fixed Frequency (PLL Lockable) Wide VIN Range: 3.5V to 36V Operation Ultrahigh Efficiency Very Low Dropout Operation: 99% Duty Cycle Low Dropout, 0.5A Linear Regulator for VPP Generation or Low Noise Audio Supply Built-In Power-On Reset Timer Programmable Soft Start Low-Battery Detector Remote Output Voltage Sense Foldback Current Limiting (Optional) Pin Selectable Output Voltage 5V Standby Regulator Active in Shutdown: IQ < 200A Output Voltages from 1.19V to 9V Available in 28- and 36-Lead SSOP Packages
APPLICATIONS
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Notebook and Palmtop Computers, PDAs Portable Instruments Battery-Operated Devices DC Power Distribution Systems
TYPICAL APPLICATION
DB1, CMDSH-3 BOOST 1 M1 L1 10H D1 MBR140T3 M3* CB1 0.1F M2 TGL1 TGS1 SW1 BG1 SENSE + 1 RSENSE1 0.03 VOUT1 5V 3.5A 1000pF CC1 1000pF CC1A 220pF SENSE - 1 ITH1 RUN/SS1 RC1 10k CSS1 0.1F CDSC VPROG2 COSC 56pF SGND VOSENSE2 ITH2 PGND RUN/SS2 CSS2 0.1F LTC1539 SENSE + 2 SENSE - 2 VPROG1 VIN INTVCC DB2, CMDSH-3 BOOST 2 TGL2 TGS2 SW2 BG2
5V STANDBY
+
CB2, 0.1F M5
1000pF
+
COUT1 220F 10V M1, M2, M4, M5: Si4412DY
CC2 1000pF RC2 10k CC2A 470pF
M3, M6: IRLML2803
*NOT REQUIRED FOR LTC1538-AUX
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 1. High Efficiency Dual 5V/3V Step-Down Converter
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4.7F 16V
+
VIN CIN 5.2V TO 28V 22F 35V x4
M4 M6* L2 10H D2 MBR140T3 RSENSE2 0.03 VOUT2 3.3V 3.5A
+
1538 F01
COUT 220F 10V
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LTC1538-AUX/LTC1539
ABSOLUTE MAXIMUM RATINGS
Input Supply Voltage (VIN)....................... 36V to - 0.3V Topside Driver Voltage (BOOST 1, 2) ...... 42V to - 0.3V Peak Switch Voltage > 10s (SW 1, 2) ... VIN + 5V to - 5V EXTVCC Voltage........................................ 10V to - 0.3V POR1, LBO Voltages ................................ 12V to - 0.3V AUXFB Voltage ........................................ 20V to - 0.3V AUXDR Voltage ........................................ 28V to - 0.3V SENSE+ 1, SENSE+ 2, SENSE- 1, SENSE- 2, VOSENSE2 Voltages ................... INTVCC + 0.3V to - 0.3V VPROG1, VPROG2 Voltages .................... INTVCC to - 0.3V PLL LPF, ITH1, ITH2 Voltages ................... 2.7V to - 0.3V AUXON, PLLIN, SFB1, RUN/SS1, RUN/SS2, LBI, Voltages ......... 10V to - 0.3V Peak Output Current < 10s (TGL1, 2, BG1, 2) ......... 2A Peak Output Current < 10s (TGS1, 2) .............. 250mA INTVCC Output Current ........................................ 50mA Operating Temperature Range LTC1538-AUXCG/LTC1539CGW............ OC to 70C LTC1538-AUXIG/LTC1539IGW .......... - 40C to 85C Junction Temperature (Note 1) ............................ 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
PACKAGE/ORDER INFORMATION
TOP VIEW BOOST 1 RUN/SS1 SENSE + 1 SENSE - 1 VPROG1 ITH1 COSC SGND SFB1 1 2 3 4 5 6 7 8 9 28 TGL1 27 SW1 26 VIN 25 BG1 24 INTVCC /5V 23 PGND 22 BG2 21 EXTVCC 20 SW2 19 TGL2 18 BOOST 2 17 AUXON 16 AUXFB 15 AUXDR
ORDER PART NUMBER LTC1538CG-AUX LTC1538IG-AUX
ITH2 10 VOSENSE2 11 SENSE - 2 12 SENSE + 2 13
RUN/SS2 14
G PACKAGE 28-LEAD PLASTIC SSOP
TJMAX = 125C, JA = 95C/ W
Consult factory for Military grade parts.
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TOP VIEW RUN/SS1 SENSE + 1 SENSE - 1 VPROG1 ITH1 POR1 COSC SGND LBI 1 2 3 4 5 6 7 8 9 36 PLL LPF 35 PLLIN 34 BOOST 1 33 TGL1 32 SW1 31 TGS1 30 VIN 29 BG1 28 INTVCC /5V 27 PGND 26 BG2 25 EXTVCC 24 TGS2 23 SW2 22 TGL2 21 BOOST 2 20 AUXON 19 AUXFB
ORDER PART NUMBER LTC1539CGW LTC1539IGW
LBO 10 SFB1 11 ITH2 12 VPROG2 13 VOSENSE2 14 SENSE - 2 15 SENSE+ 2 16
RUN/SS2 17 AUXDR 18
GW PACKAGE 36-LEAD PLASTIC SSOP TJMAX = 125C, JA = 85C/ W
LTC1538-AUX/LTC1539
ELECTRICAL CHARACTERISTICS
SYMBOL PARAMETER Main Control Loops IIN VOSENSE2 Feedback Current Regulated Output Voltage VOUT1,2 1.19V (Adjustable) Selected 3.3V Selected 5V Selected VLINEREG1,2 Reference Voltage Line Regulation VLOADREG1,2 Output Voltage Load Regulation VSFB1 ISFB1 VOVL IPROG1,2 IQ Secondary Feedback Threshold Secondary Feedback Current Output Overvoltage Lockout VPROG1,2 Input Current
TA = 25C, VIN = 15V, VRUN/SS1,2 = 5V unless otherwise noted.
MIN TYP 10
q q q q q q
CONDITIONS VPROG1, VPROG2 Pins Open (Note 2) (Note 2) VPROG1, VPROG2 Pins Open VPROG1, VPROG2 = 0V VPROG1, VPROG2 = INTVCC VIN = 3.6V to 20V (Note 2), VPROG1,2 Pins Open ITH1,2 Sinking 5A (Note 2) ITH1,2 Sourcing 5A VSFB1 Ramping Negative VSFB1 = 1.5V VPROG1,2 Pin Open, SENSE - 1 and VOSENSE2 Pins 0.5V > VPROG1,2 INTVCC - 0.5V < VPROG1,2 < INTVCC EXTVCC = 5V (Note 3) 3.6V < VIN < 30V, VAUXON = 0V VRUN/SS1,2 = 0V, 3.6V < VIN < 15V VRUN/SS1,2 = 0V VOSENSE1,2 = 0V, 5V VPROG1,2 = Pins Open CLOAD = 3000pF CLOAD = 3000pF CLOAD = 500pF CLOAD = 500pF CLOAD = 3000pF CLOAD = 3000pF 6V < VIN < 30V, VEXTVCC = 4V INTVCC = 20mA, VEXTVCC = 4V INTVCC = 20mA, VEXTVCC = 5V INTVCC = 20mA, EXTVCC Ramping Positive COSC = 100pF, LTC1539: PLL LPF = 0V (Note 4) LTC1539, VPLLLPF = 2.4V LTC1539 fPLLIN < fOSC fPLLIN > fOSC IPOR1 = 1.6mA, VOSENSE1 = 1V, VPROG1 Pins Open VPOR1 = 12V, VOSENSE1 = 1.19V, VPROG1 Pin Open VPROG1 Pin Open % of VREF VOSENSE1 Ramping Negative VPROG1 Pin Open
q
MAX 50 1.202 3.380 5.100 0.01 0.8 - 0.8 1.22 -2 1.32 -6 6
UNITS nA V V V %/V % % V A V A A A A V A mV ns ns ns ns ns ns V % mV V kHz kHz k A A V A % Cycles
1.178 3.220 4.900
1.16 1.24
1.19 3.30 5.00 0.002 0.5 - 0.5 1.19 -1 1.28 -3 3 320 70 1.3 3 150 50 50 100 50 50 50
Input DC Supply Current Normal Mode Shutdown VRUN/SS1,2 Run Pin Threshold IRUN/SS1,2 Soft Start Current Source VSENSE(MAX) Maximum Current Sense Threshold TGL1, 2 t r, t f TGL1, TGL2 Transition Time Rise Time Fall Time TGS1, 2 t r, t f TGS1, TGS2 Transition Time Rise Time Fall Time BG1, BG2 Transition Time BG1, 2 t r, t f Rise Time Fall Time Internal VCC Regulator - 5V Standby VINTVCC Internal VCC Voltage VLDO INT INTVCC Load Regulation VLDO EXT EXTVCC Voltage Drop VEXTVCC EXTVCC Switchover Voltage Oscillator and Phase-Locked Loop Oscillator Frequency fOSC VCO High RPLLIN PLLIN Input Resistance Phase Detector Output Current IPLLLPF Sinking Capability Sourcing Capability Power-On Reset POR1 Saturation Voltage VSATPOR1 ILPOR1 VTHPOR1 tDPOR1 POR1 Leakage POR1 Trip Voltage POR1 Delay
q
0.8 1.5 130
200 2 4.5 180 150 150 150 150 150 150 5.2 -1 300
4.8
q
4.5 112 200
5.0 - 0.2 170 4.7 125 240 50 15 15 0.6 0.2
138
10 10
20 20 1 1 -4
- 11
- 7.5 65536
3
LTC1538-AUX/LTC1539
ELECTRICAL CHARACTERISTICS
SYMBOL PARAMETER Low-Battery Comparator VSATLBO LBO Saturation Voltage ILLBO LBO Leakage VTHLB1 LBI Trip Voltage IINLB1 LBI Input Current VHYSLBO LBO Hysteresis Auxiliary Regulator/Comparator AUXDR Current IAUXDR Max Current Sinking Capability Control Current Leakage when OFF IINAUXFB AUXFB Input Current IINAUXON AUXON Input Current VTHAUXON AUXON Trip Voltage VSATAUXDR AUXDR Saturation Voltage VAUXFB AUXFB Voltage VTHAUXDR AUXFB Divider Disconnect Voltage
TA = 25C, VIN = 15V, VRUN/SS1,2 = 5V unless otherwise noted.
MIN TYP 0.6 0.01 1.19 1 20 MAX 1 1 1.22 50 UNITS V A V nA mV
CONDITIONS ILBO = 1.6mA, VLBI = 1.1V VLBO = 12V, VLBI = 1.4V High to Low Transition on LBO VLBI = 1.19V
q q q
1.16
VEXTVCC = 0V VAUXDR = 4V, VAUXFB = 1.0V, VAUXON = 5V VAUXDR = 5V, VAUXFB = 1.5V, VAUXON = 5V VAUXDR = 24V, VAUXFB = 1.5V, VAUXON = 0V VAUXFB = 1.19V, VAUXON = 5V VAUXON = 5V VAUXDR = 4V, VAUXFB = 1V IAUXDR = 1.6mA, VAUXFB = 1V, VAUXON = 5V VAUXON = 5V, 11V < VAUXDR < 24V (Note 5) VAUXON = 5V, 3V < VAUXDR < 7V VAUXON = 5V (Note 5); Ramping Negative
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1.0
q q
11.5 1.14 7.5
15 1 0.01 0.01 0.01 1.19 0.4 12.00 1.19 8.5
5 1 1 1 1.4 0.8 12.5 1.24 9.5
mA A A A A V V V V V
The q denotes specifications which apply over the full operating temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1538CG-AUX: TJ = TA + (PD)(95C/W) LTC1539CGW: TJ = TA + (PD)(85C/W) Note 2: The LTC1538-AUX and LTC1539 are tested in a feedback loop which servos VOSENSE1,2 to the balance point for the error amplifier (VITH1,2 = 1.19V).
Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 4: Oscillator frequency is tested by measuring the COSC charge and discharge current (IOSC) and applying the formula: fOSC (kHz) = 8.4(108)[COSC (pF) + 11]-1 (1/ICHG + 1/IDISC) -1 Note 5: The auxiliary regulator is tested in a feedback loop which servos VAUXFB to the balance point for the error amplifier. For applications with VAUXDR > 9.5V, VAUXFB uses an internal resistive divider. See Applications Information section.
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LTC1538-AUX/LTC1539 TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage: VOUT = 3.3V
100 VOUT = 3.3V 95 ILOAD = 1A EFFICIENCY (%) EFFICIENCY (%) 90 85 ILOAD = 100mA 80 75 70 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 90 ILOAD = 100mA 85 80 75 70 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 95 ILOAD = 1A
EFFICIENCY (%)
VIN - VOUT Dropout Voltage vs Load Current
0.5 0 RSENSE = 0.033 VOUT DROP OF 5%
0.4
VIN - VOUT (V)
VOUT (%)
- 0.75 -1.00
VITH (V)
0.3
0.2
0.1
0 0 0.5 1.0 1.5 2.0 LOAD CURRENT (A) 2.5 3.0
Input Supply Current vs Input Voltage
INTVCC PERCENT CHANGE, NORMALIZED (V)
2.5
2.0
SUPPLY CURRENT (mA)
5V, 3.3V OFF 5V STANDBY 1.5 60
70C
EXTVCC - INTVCC (mV)
1.0
5V, 3.3V ON 5V OFF, 3.3V ON
0.5 5V ON, 3.3V OFF 0
0
5
10 15 20 INPUT VOLTAGE (V)
LTC1538/39 * TPC07
Adaptive Power and Burst Mode are trademarks of Linear Technology Corporation.
UW
1538/39 * G01 1538/39 * G04
Efficiency vs Input Voltage: VOUT = 5V
100 VOUT = 5V
100 95 90 85 80 75 70 65 60 55
Efficiency vs Load Current
VIN = 10V VOUT = 5V RSENSE = 0.33 CONTINUOUS MODE Burst ModeTM OPERATION
Adaptive PowerTM MODE
50 0.001
1 0.01 0.1 LOAD CURRENT (A)
10
1538/39 * G03
1538/39 * G02
Load Regulation
RSENSE = 0.033 - 0.25 - 0.50
VITH Pin Voltage vs Output Current
3.0 2.5 2.0 1.5 Burst Mode OPERATION 1.0 0.5 0 0 10 20 30 40 50 60 70 80 90 100 OUTPUT CURRENT (%)
1538/39 * G06
CONTINUOUS/ Adaptive Power MODE
-1.25 -1.50 0 0.5 1.0 1.5 2.0 LOAD CURRENT (A) 2.5 3.0
1538/39 * G05
INTVCC Regulation vs INTVCC Load Current
100
2 EXTVCC = 0V 1
EXTVCC Switch Drop vs INTVCC Load Current
300
80
5V STANDBY CURRENT (A)
70C 200 25C - 45C 100
0 25C -1
40
20
0 25 30
-2
0
0 20 30 40 10 INTVCC LOAD CURRENT (mA) 50
0
5
15 20 25 10 INTVCC LOAD CURRENT (mA)
30
1538/39 * G08
1538/39 * G09
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LTC1538-AUX/LTC1539 TYPICAL PERFORMANCE CHARACTERISTICS
Normalized Oscillator Frequency vs Temperature
10 4
RUN/SS CURRENT (A)
5
SFB CURRENT (A)
FREQUENCY (%)
fO
-5
-10 - 40 -15
60 35 85 10 TEMPERATURE (C)
Maximum Current Comparator Threshold Voltage vs Temp
154
CURRENT SENSE THRESHOLD (mV)
152
150
148
146 - 40 -15
85 10 35 60 TEMPERATURE (C)
Burst Mode Operation
AUXILIARY OUTPUT VOLTAGE (V)
VOUT 20mV/DIV
VOUT 200mV/DIV
ILOAD = 50mA
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110
1538/39 * G10
RUN/SS Pin Current vs Temperature
0 - 0.25 3 -1.50 - 0.75 -1.00 -1.25 0 - 40 -15
SFB1 Pin Current vs Temperature
2
1
135
85 10 35 60 TEMPERATURE (C)
110
135
-1.50 - 40 -15
60 35 85 10 TEMPERATURE (C)
110
135
1538/39 * G11
1538/39 * G12
Transient Response
Transient Response
VOUT 50mV/DIV
VOUT 50mV/DIV
ILOAD = 50mA to 1A
1538/39 * G14
ILOAD = 1A to 3A
1538/39 * G15
110
135
1538/39 * G13
Soft Start: Load Current vs Time
12.2
Auxiliary Regulator Load Regulation
EXTERNAL PNP: 2N2907A
RUN/SS 5V/DIV
12.1
12.0
INDUCTOR CURRENT 1A/DIV
11.9
1538/39 * G16
1538/39 * G17
11.8
11.7
0
40 120 160 80 AUXILIARY LOAD CURRENT (mA)
200
1538/39 * G18
LTC1538-AUX/LTC1539 TYPICAL PERFORMANCE CHARACTERISTICS
Auxiliary Regulator Sink Current Available
20
100 90
AUX DR CURRENT (mA)
15
10
PSRR (dB)
5
0 0 2 4 10 12 6 8 AUX DR VOLTAGE (V) 14 16
PIN FUNCTIONS
VIN: Main Supply Pin. Must be closely decoupled to the IC's signal ground pin. INTVCC/5V STANDBY: Output of the Internal 5V Regulator and the EXTVCC Switch. The driver and control circuits are powered from this voltage. Must be closely decoupled to power ground with a minimum of 2.2F tantalum or electrolytic capacitor. The INTVCC regulator remains on when both RUN/SS1 and RUN/SS2 are low. Refer to the LTC1438/LTC1439 for applications which do not require a 5V standby regulator. EXTVCC: External Power Input to an Internal Switch. This switch closes and supplies INTVCC, bypassing the internal low dropout regulator whenever EXTVCC is higher than 4.8V. Connect this pin to VOUT of the controller with the higher output voltage. Do not exceed 10V on this pin. See EXTVCC connection in Applications Information section. BOOST 1, BOOST 2: Supplies to the Topside Floating Drivers. The bootstrap capacitors are returned to these pins. Voltage swing at these pins is from INTVCC to VIN + INTVCC. SW1, SW2: Switch Node Connections to Inductors. Voltage swing at these pins is from a Schottky diode (external) voltage drop below ground to VIN. SGND: Small Signal Ground. Common to both controllers, must be routed separately from high current grounds to the (-) terminals of the COUT capacitors. PGND: Driver Power Ground. Connects to sources of bottom N-channel MOSFETs and the (-) terminals of CIN. SENSE - 1, SENSE - 2: Connects to the (-) input for the current comparators. SENSE- 1 is internally connected to the first controllers VOUT sensing point preventing true remote output voltage sensing operation. The first controller can only be used as a 3.3V or 5.0V regulator controlled by the VPROG1 pin. The second controller can be set to a 3.3V, 5.0V or an adjustable regulator controlled by the VPROG2 pin (see Table 1).
Table 1. Output Voltage Table
CONTROLLER 1 CONTROLLER 2 LTC1538-AUX LTC1539 5V or 3.3V Only, Secondary Feedback Loop Adjustable Only Remote Sensing 5V/3.3V/Adjustable Remote Sensing POR1 Output
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Auxiliary Regulator PSRR
80 70 60 50 40 30 20 10 0 10
IL = 10mA IL = 100mA
100 FREQUENCY (kHz)
1000
1538/39 * G20
1538/39 * G19
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LTC1538-AUX/LTC1539
PIN FUNCTIONS
SENSE + 1, SENSE + 2: The (+) Input to Each Current Comparator. Built-in offsets between SENSE- 1 and SENSE+ 1 pins in conjunction with RSENSE1 set the current trip threshold (same for second controller). VOSENSE2: Receives the remotely sensed feedback voltage for the second controller either from the output directly or from an external resistive divider across the output . The VPROG2 pin determines which point. VOSENSE2 must connect to. VPROG1, VPROG2: Programs Internal Voltage Attenuators for Output Voltage Sensing. The voltage sensing for the first controller is internally connected to SENSE- 1 while the VOSENSE2 pin allows for remote sensing for the second controller. For VPROG1, VPROG2 < VINTVCC /3, the divider is set for an output voltage of 3.3V. With V PROG1 , VPROG2 > VINTVCC /1.5 the divider is set for an output voltage of 5V. Leaving VPROG2 open (DC) allows the output voltage of the second controller to be set by an external resistive divider connected to VOSENSE2. COSC: External capacitor COSC from this pin to ground sets the operating frequency. ITH1, ITH2: Error Amplifier Compensation Point. Each associated current comparator threshold increases with this control voltage. RUN/SS1, RUN/SS2: Combination of Soft Start and RUN Control Inputs. A capacitor to ground at each of these pins sets the ramp time to full current output. The time is approximately 0.5s/F. Forcing either of these pins below 1.3V causes the IC to shut down the circuitry required for that particular controller. Forcing both of these pins below 1.3V causes the device to shut down both controllers, leaving the 5V standby regulator, internal reference and a comparator active. Refer to the LTC1438/LTC1439 for applications which do not require a 5V standby regulator. TGL1, TGL2: High Current Gate Drives for Main Top N-Channel MOSFET. These are the outputs of floating drivers with a voltage swing equal to INTVCC superimposed on the switch node voltage SW1 and SW2. TGS1, TGS2: Gate Drives for Small Top N-Channel MOSFET. These are the outputs of floating drivers with a voltage swing equal to INTVCC superimposed on the switch node voltage SW. Leaving TGS1 or TGS2 open invokes Burst Mode operation for that controller. BG1, BG2: High Current Gate Drive Outputs for Bottom NChannel MOSFETs. Voltage swing at these pins is from ground to INTVCC. SFB1: Secondary Winding Feedback Input. This input acts only on the first controller and is normally connected to a feedback resistive divider from the secondary winding. Pulling this pin below 1.19V will force continuous synchronous operation for the first controller. This pin should be tied to: ground to force continuous operation; INTVCC in applications that don't use a secondary winding; and a resistive divider from the output in applications using a secondary winding. POR1: This output is a drain of an N-channel pull-down. This pin sinks current when the output voltage of the first controller drops 7.5% below its regulated voltage and releases 65536 oscillator cycles after the output voltage of the first controller rises to within -5% value of its regulated value. The POR1 output is asserted when RUN/SS1 and RUN/SS2 are both low, independent of the VOUT1. LBO: This output is a drain of an N-channel pull-down. This pin will sink current when the LBI pin goes below 1.19V irrespective of the RUN/SS pin voltage. LBI: The (+) input of a comparator which can be used as a low-battery voltage detector irrespective of the RUN/SS pin voltage. The (-) input is connected to the 1.19V internal reference. PLLIN: External Synchronizing Input to Phase Detector. This pin is internally terminated to SGND with 50k. Tie this pin to SGND in applications which do not use the phase-locked loop. PLL LPF: Output of Phase Detector and Control Input of Oscillator. Normally a series RC lowpass filter network is connected from this pin to ground. Tie this pin to SGND in applications which do not use the phase-locked loop. Can be driven by a 0V to 2.4V logic signal for a frequency shifting option. AUXFB: Feedback Input to the Auxiliary Regulator/Comparator. When used as a linear regulator, this input can either be connected to an external resistive divider or directly to the collector of the external PNP pass device for 12V operation. When used as a comparator, this is the
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LTC1538-AUX/LTC1539
PIN FUNCTIONS
noninverting input of a comparator whose inverting input is tied to the internal 1.19V reference. See Auxiliary Regulator Application section. AUXON: Pulling this pin high turns on the auxiliary regulator/comparator. The threshold is 1.19V. This is a convenient linear power supply logic-controlled on/off input. AUXDR: Open Drain Output of the Auxiliary Regulator/ Comparator. The base of an external PNP device is connected to this pin when used as a linear regulator. An external pull-up resistor is required for use as a comparator. A voltage > 9.5V on AUXDR causes the internal 12V resistive divider to be connected in series with the AUXFB pin.
FUNCTIONAL DIAGRA
PLLIN** fIN 50k RLP PLL LPF** CLP COSC COSC OSCILLATOR POR1** POWER-ON RESET VFB1 1.11V PHASE DETECTOR 2.4V
LBI** BATTERY SENSE LBO**
I1
+ + - -
I2 30k 8k INTVCC
9V
-
+
AUXDR VLDO AUXFB
SENSE - SENSE + VOSENSE*
4k 10k 180k VFB 320k 61k
90.8k
+
EA SFB 1A 1.19V REF gm = 1m VREF OV
+ +
SFB1*
-
119k
-
+ -
DFB
VIN VIN
4.8V
+
5V LDO REGULATOR 3A 1.28V 1.19V SHUTDOWN RUN SOFT START ITH CC
EXTVCC
-
RC
INTVCC
6V
RUN/SS CSS
+
SGND INTERNAL SUPPLY
*NOT AVAILABLE ON BOTH CHANNELS
**NOT AVAILABLE ON LTC1538-AUX
FOLDBACK CURRENT LIMITING OPTION
BOLD LINES INDICATE HIGH CURRENT PATHS
1438 FD
9
+
VPROG*
+
VSEC
AUXON
-
+
W
-
-
+
+
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LTC1539 shown, see specific package pinout for availability of specific functions.
DUPLICATE FOR SECOND CONTROLLER CHANNEL
BOOST
VIN
INTVCC DB
TGL DROPOUT DETECTOR SFB SQ RQ 0.6V TGS** CB
+ -
SHUTDOWN
SWITCH LOGIC INTVCC
SW
*
BG
+
PGND COUT
*
CIN
RSENSE
VOUT
CSEC
LTC1538-AUX/LTC1539
OPERATION
Main Control Loop The LTC1538-AUX/LTC1539 use a constant frequency, current mode step-down architecture. During normal operation, the top MOSFET is turned on each cycle when the oscillator sets the RS latch and turned off when the main current comparator I1 resets the RS latch. The peak inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH1 (ITH2) pin, which is the output of each error amplifier (EA). The VPROG1 pin, described in the Pin Functions, allows the EA to receive a selectively attenuated output feedback voltage VFB1 from the SENSE- 1 pin while VPROG2 and VOSENSE2 allow EA to receive an output feedback voltage VFB2 from either internal or external resistive dividers on the second controller. When the load current increases, it causes a slight decrease in VFB relative to the 1.19V reference, which in turn causes the ITH1 (ITH2) voltage to increase until the average inductor current matches the new load current. After the large top MOSFET has turned off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle. The top MOSFET drivers are biased from floating boot strap capacitor CB, which normally is recharged during each Off cycle. When VIN decreases to a voltage close to VOUT, however, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector counts the number of oscillator cycles that the top MOSFET remains on and periodically forces a brief off period to allow CB to recharge. The main control loop is shut down by pulling the RUN/ SS1 (RUN/SS2) pin low. Releasing RUN/SS1 (RUN/SS2) allows an internal 3A current source to charge soft start capacitor CSS. When CSS reaches 1.3V, the main control loop is enabled with the ITH1 (ITH2) voltage clamped at approximately 30% of its maximum value. As CSS continues to charge, ITH1 (ITH2) is gradually released allowing normal operation to resume. When both RUN/SS1 and RUN/SS2 are low, all LTC1538-AUX/LTC1539 functions are shut down except for the 5V standby regulator, internal reference and a comparator. Refer to the LTC1438/LTC1439 for applications which do not require a 5V standby regulator.
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(Refer to Functional Diagram)
Comparator OV guards against transient overshoots > 7.5% by turning off the top MOSFET and keeping it off until the fault is removed. Low Current Operation Adaptive Power Mode allows the LTC1539 to automatically change between two output stages sized for different load currents. The TGL1 (TGL2) and BG1 (BG2) pins drive large synchronous N-channel MOSFETs for operation at high currents, while the TGS1 (TGS2) pin drives a much smaller N-channel MOSFET used in conjunction with a Schottky diode for operation at low currents. This allows the loop to continue to operate at normal operating frequency as the load current decreases without incurring the large MOSFET gate charge losses. If the TGS1 (TGS2) pin is left open, the loop defaults to Burst Mode operation in which the large MOSFETs operate intermittently based on load demand. Adaptive Power mode provides constant frequency operation down to approximately 1% of rated load current. This results in an order of magnitude reduction of load current before Burst Mode operation commences. Without the small MOSFET (ie: no Adaptive Power mode) the transition to Burst Mode operation is approximately 10% of rated load current. The transition to low current operation begins when comparator I2 detects current reversal and turns off the bottom MOSFET. If the voltage across RSENSE does not exceed the hysteresis of I2 (approximately 20mV) for one full cycle, then on following cycles the top drive is routed to the small MOSFET at the TGS1 (TGS2) pin and the BG1 (BG2) pin is disabled. This continues until an inductor current peak exceeds 20mV/RSENSE or the ITH1 (ITH2) voltage exceeds 0.6V, either of which causes drive to be returned to the TGL1 (TGL2) pin on the next cycle. Two conditions can force continuous synchronous operation, even when the load current would otherwise dictate low current operation. One is when the common mode voltage of the SENSE+ 1 (SENSE+ 2) and SENSE- 1 (SENSE- 2) pins are below 1.4V, and the other is when the SFB1 pin is below 1.19V. The latter condition is used to assist in secondary winding regulation, as described in the Applications Information section.
LTC1538-AUX/LTC1539
OPERATION
Frequency Synchronization A Phase-Locked Loop (PLL) is available on the LTC1539 to allow the oscillator to be synchronized to an external source connected to the PLLIN pin. The output of the phase detector at the PLL LPF pin is also the control input of the oscillator, which operates over a 0V to 2.4V range corresponding to - 30% to 30% in frequency. When locked, the PLL aligns the turn-on of the top MOSFET to the rising edge of the synchronizing signal. When PLLIN is left open, PLL LPF goes low, forcing the oscillator to minimum frequency. Power-On Reset The POR1 pin is an open drain output which pulls low when the main regulator output voltage of the LTC1539 first controller is out of regulation. When the output voltage rises to within 5% of regulation, a timer is started which releases POR1 after 216 (65536) oscillator cycles. Auxiliary Linear Regulator The auxiliary linear regulator in the LTC1538-AUX and LTC1539 controls an external PNP transistor for operation up to 500mA. A precise internal AUXFB resistive divider is invoked when the AUXDR pin is above 9.5V to allow regulated 12V VPP supplies to be easily implemented. When AUXDR is below 8.5V an external feedback divider may be used to set other output voltages. Taking the AUXON pin low shuts down the auxiliary regulator providing a convenient logic-controlled power supply.
APPLICATIONS INFORMATION
The basic LTC1539 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, COSC and L can be chosen. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by the external MOSFETs). RSENSE Selection for Output Current RSENSE is chosen based on the required output current. The LTC1538-AUX/LTC1539 current comparator has a maximum threshold of 150mV/RSENSE and an input common mode range of SGND to INTVCC. The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current, IL.
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(Refer to Functional Diagram)
The AUX block can be used as a comparator having its inverting input tied to the internal 1.19V reference. The AUXDR pin is used as the output and requires an external pull-up to a supply of less than 8.5V in order to inhibit the invoking of the internal resistive divider. INTVCC / EXTVCC Power Power for the top and bottom MOSFET drivers and most of the other LTC1538-AUX/LTC1539 circuitry is derived from the INTVCC pin. The bottom MOSFET driver supply is also connected to INTVCC. When the EXTVCC pin is left open, an internal 5V low dropout regulator supplies INTVCC power. If EXTVCC is taken above 4.8V, the 5V regulator is turned off and an internal switch is turned on to connect EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source such as the output of the regulator itself or a secondary winding, as described in the Applications Information section. The 5V/20mA INTVCC regulator can be used as a standby regulator when the two controllers are in shutdown or when either or both controllers are on. Irrespective of the signals on the RUN/SS pins, the INTVCC pin will follow the voltage applied to the EXTVCC pin when the voltage applied to the EXTVCC pin is taken above 4.8V. The externally applied voltage is required to be less than the voltage applied to the VIN pin at all times, even when both controllers are shut down. This prevents a voltage backfeed situation from the source applied to the EXTVCC pin to the VIN pin. If the EXTVCC pin is tied to the first controller's 5V output, the nominal INTVCC pin voltage will stay in the guaranteed range of 4.7V to 5.2V.
11
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
Allowing some margin for variations in the LTC1538-AUX/ LTC1539 and external component values yield:
100mV IMAX The LTC1538-AUX/LTC1539 work well with values of RSENSE from 0.005 to 0.2. RSENSE =
COSC Selection for Operating Frequency The LTC1538-AUX/LTC1539 use a constant frequency architecture with the frequency determined by an external oscillator capacitor on COSC. Each time the topside MOSFET turns on, the voltage on COSC is reset to ground. During the on-time, COSC is charged by a fixed current plus an additional current which is proportional to the output voltage of the phase detector (VPLLLPF)(LTC1539 only). When the voltage on the capacitor reaches 1.19V, COSC is reset to ground. The process then repeats. The value of COSC is calculated from the desired operating frequency. Assuming the phase-locked loop has no external oscillator input (VPLLLPF = 0V):
C OSC (pF ) = 1.37 104 Frequency (kHz )
()
- 11
A graph for selecting COSC vs frequency is given in Figure 2. As the operating frequency is increased the gate charge losses will be higher, reducing efficiency (see Efficiency Considerations). The maximum recommended switching frequency is 400kHz. When using Figure 2 for
300 VPLLLPF = 0V 250
INDUCTOR VALUE (H)
COSC VALUE (pF)
200 150 100 50 0
0
100 200 300 400 OPERATING FREQUENCY (kHz)
500
LTC1538 * F02
Figure 2. Timing Capacitor Value
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synchronizable applications, choose COSC corresponding to a frequency approximately 30% below your center frequency. (See Phase-Locked Loop and Frequency Sychronization). Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic trade off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current IL decreases with higher inductance or frequency and generally increases with higher VIN or VOUT: IL = V 1 VOUT 1 - OUT (f)(L) VIN
Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IL = 0.4(IMAX). Remember, the maximum IL occurs at the maximum input voltage. The inductor value also has an effect on low current operation. The transition to low current operation begins
60 50 40 30 20 10 0 VOUT = 5.0V VOUT = 3.3V VOUT = 2.5V
0
100 150 200 250 50 OPERATING FREQUENCY (kHz)
300
1538 F03
Figure 3. Recommended Inductor Values
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
when the inductor current reaches zero while the bottom MOSFET is on. Lower inductor values (higher IL) will cause this to occur at higher load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation (TGS1, 2 pins open), lower inductance values will cause the burst frequency to decrease. The Figure 3 graph gives a range of recommended inductor values vs operating frequency and VOUT. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool M(R) cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool M. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available which do not increase the height significantly. Power MOSFET and D1 Selection Three external power MOSFETs must be selected for each controller with the LTC1539: a pair of N-channel MOSFETs for the top (main) switch and an N-channel MOSFET for the bottom (synchronous) switch. Only one top MOSFET is required for each LTC1538-AUX controller.
Kool M is a registered trademark of Magnetics, Inc.
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To take advantage of the Adaptive Power output stage, two topside MOSFETs must be selected. A large [low RSD(ON)] MOSFET and a small [higher RDS(ON)] MOSFET are required. The large MOSFET is used as the main switch and works in conjunction with the synchronous switch. The smaller MOSFET is only enabled under low load current conditions. The benefit of this is to boost low to midcurrent efficiencies while continuing to operate at constant frequency. Also, by using the small MOSFET the circuit will keep switching at a constant frequency down to lower currents and delay skipping cycles. The RDS(ON) recommended for the small MOSFET is around 0.5. Be careful not to use a MOSFET with an RDS(ON) that is too low; remember, we want to conserve gate charge. (A higher RDS(ON) MOSFET has a smaller gate capacitance and thus requires less current to charge its gate). For all LTC1538-AUX and cost sensitive LTC1539 applications, the small MOSFET is not required. The circuit then begins Burst Mode operation as the load current drops. The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see EXTVCC Pin Connection). Consequently, logic level threshold MOSFETs must be used in most LTC1538-AUX/ LTC1539 applications. The only exception is applications in which EXTVCC is powered from an external supply greater than 8V (must be less than 10V), in which standard threshold MOSFETs (VGS(TH) < 4V) may be used. Pay close attention to the BVDSS specification for the MOSFETs as well; many of the logic level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the "ON" resistance RSD(ON), reverse transfer capacitance CRSS, input voltage and maximum output current. When the LTC1538-AUX/LTC1539 are operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by:
V Main Switch Duty Cycle = OUT VIN Synchronous Switch Duty Cycle =
(VIN - VOUT)
VIN
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
The MOSFET power dissipations at maximum output current are given by: V 2 PMAIN = OUT (IMAX ) (1 + )RDS(ON) + VIN k (VIN)
1.85
(IMAX)(CRSS )( f)
V -V 2 PSYNC = IN OUT (IMAX ) (1 + ) RDS(ON) VIN where is the temperature dependency of RDS(ON) and k is a constant inversely related to the gate drive current. losses while the topside Both MOSFETs have N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CRSS actual provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage or during a short circuit when the duty cycle in this switch is nearly 100%. Refer to the Foldback Current Limiting section for further applications information. The term (1 + ) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but = 0.005/C can be used as an approximation for low voltage MOSFETs. CRSS is usually specified in the MOSFET characteristics. The constant k = 2.5 can be used to estimate the contributions of the two terms in the main switch dissipation equation. The Schottky diode D1 shown in Figure 1 serves two purposes. During continuous synchronous operation, D1 conducts during the dead-time between the conduction of the two large power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on and storing charge during the dead-time, which could cost as much as 1% in efficiency. During low current operation, D1 operates in conjunction with the small top MOSFET to provide an efficient low current output stage. A 1A Schottky is I2R
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generally a good compromise for both regions of operation due to the relatively small average current. CIN and COUT Selection In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/ VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
CIN Required IRMS IMAX
[VOUT (VIN - VOUT)]1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied the capacitance is adequate for filtering. The output ripple (VOUT) is approximated by: 1 VOUT IL ESR + 4 fC OUT where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. With IL = 0.4IOUT(MAX) the output ripple will be less than 100mV at max VIN assuming: COUT Required ESR < 2RSENSE Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest (ESR size) product of any aluminum electrolytic at a somewhat
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
higher price. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. INTVCC / 5V Standby Regulator An internal P-channel low dropout regulator produces 5V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the LTC1538-AUX/ LTC1539, as well as any "wake-up" circuitry tied externally to the INTVCC pin. The INTVCC pin regulator can supply 40mA and must be bypassed to ground with a minimum of 2.2F tantalum or low ESR electrolytic capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. To prevent any interaction due to the high transient gate currents being drawn from the external capacitor an additional series filter of 10 and 10F to SGND can be added. High input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the LTC1538-AUX/ LTC1539 to be exceeded. The IC supply current is dominated by the gate charge supply current when not using an output derived EXTVCC source. The gate charge is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 1 of the Electrical Characteristics. For example, the LTC1539 is limited to less than 21mA from a 30V supply: TJ = 70C + (21mA)(30V)(85C/W) = 124C To prevent maximum junction temperature from being exceeded, the input supply current must be checked while operating in continuous mode at maximum VIN. EXTVCC Connection The LTC1538-AUX/LTC1539 contain an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. When the voltage applied to EXTVCC rises above 4.8V, the internal regulator is turned off and an internal switch closes, connecting the EXTVCC pin to the INTVCC pin thereby supplying internal power to the IC. The switch remains closed as long as the voltage applied to EXTVCC remains above 4.5V. This allows the MOSFET driver and control power to be derived from the output during normal operation (4.8V < VOUT < 9V) and from the internal regulator when the output is out of regulation (start-up, short circuit). Do not apply greater than 10V to the EXTVCC pin and ensure that EXTVCC VIN. Significant efficiency gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of Duty Cycle/Efficiency. For 5V regulators this supply means connecting the EXTVCC pin directly to VOUT. However, for 3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the output. The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC left open (or grounded). This will cause INTVCC to be powered from the internal 5V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 2. EXTVCC connected directly to VOUT. This is the normal connection for a 5V regulator and provides the highest efficiency. 3. EXTVCC connected to an output-derived boost network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage which has been boosted to
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
LTC1538-AUX LTC1539* VIN EXTVCC R6 SFB1 R5 SGND TGL1 TGS1* SW1 BG1 PGND
1538 F04a
+
CIN
VIN L1 1:1
1N4148
VSEC
N-CH N-CH
*
RSENSE
*
N-CH
OPTIONAL EXTVCC CONNECTION 5V VSEC 9V
*TGS1 ONLY AVAILABLE ON THE LTC1539
Figure 4a. Secondary Output Loop and EXTVCC Connection
greater than 4.8V. This can be done with either the inductive boost winding as shown in Figure 4a or the capacitive charge pump shown in Figure 4b. The charge pump has the advantage of simple magnetics. 4. EXTVCC connected to an external supply. If an external supply is available in the 5V to 10V range (EXTVCC VIN) it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements. When driving standard threshold MOSFETs, the external supply must be always present during operation to prevent MOSFET failure due to insufficient gate drive. Note: care must be taken when using the connections in items 3 or 4. These connections will effect the INTVCC voltage when either or both controllers are on. Topside MOSFET Driver Supply (CB,DB) External bootstrap capacitors CB connected to the BOOST 1 and BOOST 2 pins supply the gate drive voltages for the topside MOSFETs. Capacitor CB in the Functional Diagram is charged through diode DB from INTVCC when the SW1(SW2) pin is low. When one of the topside MOSFETs is to be turned on, the driver places the CB voltage across the gate source of the desired MOSFET. This enhances the MOSFET and turns on the topside switch. The switch node voltage SW1(SW2) rises to VIN and the BOOST 1(BOOST 2) pin follows. With the topside MOSFET on, the boost voltage is above the input supply: VBOOST = VIN + VINTVCC. The value of the boost capacitor CB needs to be 100 times that of the total input capacitance of the topside MOSFET(s). The reverse breakdown on DB must be greater than VIN(MAX).
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1F LTC1538-AUX LTC1539*
+
CIN
VIN
BAT85
0.22F
BAT85
1F VOUT
EXTVCC
VIN TGL1 TGS1* N-CH N-CH L1 VN2222LL RSENSE VOUT BAT85
COUT
SW1 BG1 PGND N-CH
+
COUT
1538 F04b
*TGS1 ONLY AVAILABLE ON THE LTC1539
Figure 4b. Capacitive Charge Pump for EXTVCC
Output Voltage Programming The LTC1538-AUX/LTC1539 have pin selectable output voltage programming. The output voltage is selected by the VPROG1 (VPROG2) pin as follows: VPROG1,2 = 0V VPROG1,2 = INTVCC VPROG2 = Open (DC) VOUT1,2 = 3.3V VOUT1,2 = 5V VOUT2 = Adjustable
The top of an internal resistive divider is connected to SENSE- 1 pin in Controller 1. For fixed output voltage applications the SENSE- 1 pin is connected to the output voltage as shown in Figure 5a. When using an external resistive divider for Controller 2, the VPROG2 pin is left open
VPROG1 SENSE 1 LTC1538-AUX LTC1539 SGND
1538 F05a
-
GND: VOUT = 3.3V INTVCC: VOUT = 5V
+
VOUT COUT
Figure 5a. LTC1538-AUX/LTC1539 Fixed Output Applications
1.19V VOUT 9V VPROG2* VOSENSE2 LTC1538-AUX LTC1539 SGND 100pF R1 OPEN (DC) R2
*LTC1539 ONLY
VOUT = 1.19V 1 +
()
R2 R1
1538 F05b
Figure 5b. LTC1538-AUX/LTC1539 Adjustable Applications
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
(DC) and the VOSENSE2 pin is connected to the feedback resistors as shown in Figure 5b. Controller 2 will force the externally attenuated output voltage to 1.19V. Power-On Reset Function (POR) The power-on reset function monitors the output voltage of the first controller and turns on an open drain device when it is below its properly regulated voltage. An external pull-up resistor is required on the POR1 pin. When power is first applied or when coming out of shutdown, the POR1 output is held at ground. When the output voltage rises above a level which is 5% below the final regulated output value, an internal counter starts. After this counter counts 216 (65536) clock cycles, the POR1 pulldown device turns off. The POR1 output is active when both controllers are shut down as long as VIN is powered. The POR1 output will go low whenever the output voltage of the first controller drops below 7.5% of its regulated value for longer than approximately 30s, signaling an out-of-regulation condition. In shutdown, when RUN/SS1 and RUN/SS2 are both below 1.3V, the POR1 output is pulled low even if the regulator's output is held up by an external source. RUN/ Soft Start Function The RUN/SS1 and RUN/SS2 pins each serve two functions. Each pin provides the soft start function and a means to shut down each controller. Soft start reduces surge currents from VIN by providing a gradual ramp-up of the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 3A current source charges up an external capacitor CSS. When the voltage on RUN/SS1 (RUN/SS2) reaches 1.3V the particular controller is permitted to start operating. As the voltage on the pin continues to ramp from 1.3V to 2.4V, the internal current limit is also ramped at a proportional linear rate. The current limit begins at approximately 50mV/RSENSE (at VRUN/SS = 1.3V) and ends at 150mV/RSENSE (VRUN/SS 2.7V). The output current thus ramps up slowly, reducing the starting surge current required from the input power supply. If RUN/SS has been pulled all the way to ground there is a delay before starting of approximately 500ms/F, followed by a similar time to reach full current on that controller. By pulling both RUN/SS controller pins below 1.3V, the LTC1538-AUX/LTC1539 are put into shutdown (IQ < 200A). These pins can be driven directly from logic as shown in Figure 6. Diode D1 in Figure 6 reduces the start
3.3V OR 5V D1 RUN/SS1 (RUN/SS2) RUN/SS1 (RUN/SS2)
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CSS
CSS
1538 F06
Figure 6. RUN/SS Pin Interfacing
delay but allows CSS to ramp up slowly providing the soft start function; this diode and CSS can be deleted if soft start is not needed. Each RUN/SS pin has an internal 6V Zener clamp (See Functional Diagram). Foldback Current Limiting As described in Power MOSFET and D1 Selection, the worst-case dissipation for either MOSFET occurs with a short-circuited output, when the synchronous MOSFET conducts the current limit value almost continuously. In most applications this will not cause excessive heating, even for extended fault intervals. However, when heat sinking is at a premium or higher RDS(ON) MOSFETs are being used, foldback current limiting should be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding diode DFB between the output and the ITH pin as shown in the Functional Diagram. In a hard short (VOUT = 0V) the current will be reduced to approximately 25% of the maximum output current. This technique may be used for all applications with regulated output voltages of 1.8V or greater.
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
EXTERNAL FREQUENCY 2.4V PHASE DETECTOR* *PLLIN fO DIGITAL PHASE/ FREQUENCY DETECTOR RLP CLP COSC
NORMALIZED FREQUENCY
1.3
0.7
0
0.5
1.0 1.5 VPLLLPF (V)
2.0
2.5
1538 F07
Figure 7. Operating Frequency vs VPLLLPF
Phase-Locked Loop and Frequency Synchronization The LTC1539 has an internal voltage-controlled oscillator and phase detector comprising a phase-locked loop. This allows the top MOSFET turn-on to be locked to the rising edge of an external source. The frequency range of the voltage-controlled oscillator is 30% around the center frequency fO. The value of COSC is calculated from the desired operating frequency (fO). Assuming the phase-locked loop is locked (VPLLLPF = 1.19V):
C OSC (pF ) =
2.1 10
()
4
Frequency (kHz)
- 11
Stating the frequency as a function of VPLLLPF and COSC:
Frequency (kHz) =
8.4 108
[
1 C OSC (pF ) + 11 + 2000 17 A + 18A VPLLLPF 2.4V
()
]
The phase detector used is an edge sensitive digital type which provides zero degrees phase shift between the
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*PLL LPF
COSC
OSC
SGND 50k
*LTC1539 ONLY
1538 F08
Figure 8. Phase-Locked Loop Block Diagram
external and internal oscillators. This type of phase detector will not lock up on input frequencies close to the harmonics of the VCO center frequency. The PLL hold-in range, fH, is equal to the capture range, fC: fH = fC = 0.3 fO. The output of the phase detector is a complementary pair of current sources charging or discharging the external filter network on the PLL LPF pin. A simplified block diagram is shown in Figure 8. If the external frequency fPLLIN is greater than the oscillator frequency fOSC, current is sourced continuously, pulling up the PLL LPF pin. When the external frequency is less than f0SC, current is sunk continuously, pulling down the PLL LPF pin. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. Thus the voltage on the PLL LPF pin is adjusted until the phase and frequency of the external and internal oscillators are identical. At this stable operating point the phase comparator output is open and the filter capacitor CLP holds the voltage. The LTC1539 PLLIN pin must be driven from a low impedance such as a logic gate located close to the pin. Any external attenuator used needs to be referenced to SGND. The loop filter components CLP, RLP smooth out the current pulses from the phase detector and provide a
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
stable input to the voltage-controlled oscillator. The filter components CLP and RLP determine how fast the loop acquires lock. Typically, RLP = 10k and CLP is 0.01F to 0.1F. The low side of the filter needs to be connected to SGND. The PLL LPF pin can be driven with external logic to obtain a 1:1.9 frequency shift. The circuit shown in Figure 9 will provide a frequency shift from fO to 1.9fO as the voltage on VPLLLPF increases from 0V to 2.4V. Do not exceed 2.4V on VPLLLPF.
2.4V MAX PLL LPF 18k
LTC1538 * F09
3.3V OR 5V
Figure 9. Directly Driving PLL LPF Pin
Low Battery Comparator The LTC1539 has an on-chip low battery comparator which can be used to sense a low battery condition when implemented as shown in Figure 10. This comparator is active during shutdown allowing battery charge level interrogation prior to and after powering up part or all of the system. The resistor divider R3/R4 sets the comparator trip point as follows:
R4 VLBITRIP = 1.19V 1 + R3
The divided down voltage at the negative (-) input to the comparator is compared to an internal 1.19V reference. A 20mV hysteresis is built in to assure rapid switching. The output is an open drain MOSFET and requires a pull-up resistor. This comparator is active when both the RUN/ SS1 and RUN/SS2 pins are low. The low side of the resistive divider needs to be connected to SGND.
VIN R4 LBI R3 SGND LTC1539 LBO
- +
1.19V REFERENCE
1538 F10
Figure 10. Low Battery Comparator
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SFB1 Pin Operation When the SFB1 pin drops below its ground referenced 1.19V threshold, continuous mode operation is forced. In continuous mode, the large N-channel main and synchronous switches are used regardless of the load on the main output. In addition to providing a logic input to force continuous synchronous operation, the SFB1 pin provides a means to regulate a flyback winding output. The use of a synchronous switch removes the requirement that power must be drawn from the inductor primary in order to extract power from the auxiliary winding. With the loop in continuous mode, the auxiliary output may be loaded without regard to the primary output load. The SFB1 pin provides a way to force continuous synchronous operation as needed by the flyback winding. The secondary output voltage is set by the turns ratio of the transformer in conjunction with a pair of external resistors returned to the SFB1 pin as shown in Figure 4a. The secondary regulated voltage VSEC in Figure 4a is given by:
R6 VSEC (N + 1)VOUT > 1.19V 1 + R5 where N is the turns ratio of the transformer, and VOUT is the main output voltage sensed by SENSE- 1.
Auxiliary Regulator/Comparator The auxiliary regulator/comparator can be used as a comparator or low dropout regulator (by adding an external PNP pass device). When the voltage present at the AUXON pin is greater than 1.19V the regulator/comparator is on. The amplifier is stable when operating as a low dropout regulator. This same amplifier can be used as a comparator whose inverting input is tied to the 1.19V reference. The AUXDR pin is internally connected to an open drain MOSFET which can sink up to 10mA. The voltage on AUXDR determines whether or not an internal 12V resistive divider is connected to AUXFB as described below. A pull-up resistor is required on AUXDR and the voltage must not exceed 28V.
19
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
With the addition of an external PNP pass device, a linear regulator capable of supplying up to 0.5A is created. As shown in Figure 11a, the base of the external PNP connects to the AUXDR pin together with a pull-up resistor. The output voltage VOAUX at the collector of the external PNP is sensed by the AUXFB pin. The input voltage to the auxiliary regulator can be taken from a secondary winding on the primary inductor as shown in Figure 11a. In this application, the SFB1 pin regulates the input voltage to the PNP regulator (see SFB1 Pin Operation) and should be set to approximately 1V to 2V above the required output voltage of the auxiliary regulator. A Zener clamp diode may be required to keep the secondary winding resultant output voltage under the 28V AUXDR pin specification when the primary is heavily loaded and the secondary is not. The AUXFB pin is the feedback point of the regulator. An internal resistor divider is available to provide a 12V output by simply connecting AUXFB directly to the collector of the external PNP. The internal resistive divider is switched in when the voltage at AUXFB goes above 9.5V with 1V builtSECONDARY WINDING 1:N
VSEC = 1.19V 1 +
()
R6 > 13V R5
VSEC
+
R6 SFB1
AUXDR AUXFB
R5 LTC1538-AUX/ LTC1539 AUXON
+
ON/OFF
VOAUX 12V 10F
1538 F11a
1538 F11b
Figure 11a. 12V Output Auxiliary Regulator Using Internal Feedback Resistors
ON/OFF INPUT
AUXON AUXFB
Figure 11c. Auxiliary Comparator Configuration
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in hysteresis. For other output voltages, an external resistive divider is fed back to AUXFB as shown in Figure 11b. The output voltage VOAUX is set as follows:
R8 VOAUX = 1.19V 1 + < 8V AUX DR < 8.5V R7 VOAUX = 12V AUX DR 12V
When used as a voltage comparator as shown in Figure 11c, the auxiliary block has a noninverting characteristic. When AUXFB drops below 1.19V, the AUXDR pin will be pulled low. A minimum current of 5A is required to pull up the AUXDR pin to 5V when used as a comparator output in order to counteract a 1.5A internal pull-down current source. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 + L3 + ...)
SECONDARY WINDING 1:N
VSEC = 1.19V 1 +
()
R6 > (VOAUX + 1V) R5
VSEC VOAUX
+
R6 SFB1
AUXDR AUXFB
R8
R5 LTC1538-AUX/ LTC1539 AUXON
+
R7 ON/OFF 10F
Figure 11b. 5V Output Auxiliary Regulator Using External Feedback Resistors
VPULL-UP < 7.5V LTC1538-AUX/LTC1539 AUXDR
- +
1.19V REFERENCE
OUTPUT
1538 F11c
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC1538-AUX/LTC1539 circuits. LTC1538-AUX/ LTC1539 VIN current, INTVCC current, I2R losses and topside MOSFET transition losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current typically results in a small (<< 1%) loss which increases with VIN. 2. INTVCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC which is typically much larger than the control circuit current. In continuous mode, IGATECHG = f(QT + QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. It is for this reason that the large topside and synchronous MOSFETs are turned off during low current operation in favor of the small topside MOSFET and external Schottky diode, allowing efficient, constant-frequency operation at low output currents. By powering EXTVCC from an output-derived source, the additional VIN current resulting from the driver and control currents will be scaled by a factor of Duty Cycle/ Efficiency. For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 3mA of VIN current. This reduces the midcurrent loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the MOSFET, inductor and current sense R. In continuous mode the average output current flows through L and RSENSE, but is "chopped" between the topside main MOSFET and the synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and RSENSE to obtain I2R losses. For example, if each RDS(ON) = 0.05, RL = 0.15 and RSENSE = 0.05, then the total resistance is 0.25. This results in losses ranging from 3% to 10% as the output current increases from 0.5A to 2A. I2R losses cause the efficiency to roll off at high output currents. 4. Transition losses apply only to the topside MOSFET(s) and only when operating at high input voltages (typically 20V or greater). Transition losses can be estimated from: Transition Loss 2.5(VIN)1.85(IMAX)(CRSS)(f) Other losses including CIN and COUT ESR dissipative losses, Schottky conduction losses during dead-time, and inductor core losses, generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to (ILOAD)(ESR) where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating the feedback error signal which forces the regulator loop to adapt to the current change and return VOUT to its steady-state value. During this recovery time VOUT can be monitored for overshoot or ringing which would indicate a stability problem. The ITH external components shown in Figure 1 will prove adequate compensation for most applications. A second, more severe transient is caused by switching in loads with large (> 1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25)(CLOAD). Thus a 10F capacitor would require a 250s rise time, limiting the charging current to about 200mA.
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
Automotive Considerations: Plugging into the Cigarette Lighter As battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main battery line in an automobile is the source of a number of nasty potential transients, including load dump, reverse battery and double battery. Load dump is the result of a loose battery cable. When the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes several hundred milliseconds to decay. Reverse battery is just what it says, while double battery is a consequence of tow-truck operators finding that a 24V jump start cranks cold engines faster than 12V. The network shown in Figure 12 is the most straightforward approach to protect a DC/DC converter from the ravages of an automotive battery line. The series diode prevents current from flowing during reverse battery, while the transient suppressor clamps the input voltage during load dump. Note that the transient suppressor should not conduct during double battery operation, but must still clamp the input voltage below breakdown of the converter. Although the LTC1538-AUX/LTC1539 has a maximum input voltage of 36V, most applications will be limited to 30V by the MOSFET BVDSS. Design Example As a design example, assume VIN = 12V(nominal), VIN = 22V(max), VOUT = 3.3V, IMAX = 3A and f = 250kHz, RSENSE and COSC can immediately be calculated: RSENSE = 100mV/3A = 0.033 COSC = (1.37(104 )/250) - 11 43pF Referring to Figure 3, a 10H inductor falls within the recommended range. To check the actual value of the ripple current the following equation is used : V V IL = OUT 1 - OUT (f)(L) VIN The highest value of the ripple current occurs at the maximum input voltage:
12V
50A IPK RATING VIN TRANSIENT VOLTAGE SUPPRESSOR GENERAL INSTRUMENT 1.5KA24A LTC1538-AUX/ LTC1539
Figure 12. Automotive Application Protection
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IL =
3.3V 3.3V 1- = 1.12A 250kHz(10H) 22V
The power dissipation on the topside MOSFET can be easily estimated. Using a Siliconix Si4412DY for example; RDS(ON) = 0.042, CRSS = 100pF. At maximum input voltage with T(estimated) = 50C:
PMAIN = 3.3V 2 (3) 1+ (0.005)(50C - 25C ) (0.042) 22V
1.85
[
]
+ 2.5(22V )
(3A)(100pF )(250kHz) = 122mW
The most stringent requirement for the synchronous N-channel MOSFET is with VOUT = 0V (i.e. short circuit). During a continuous short circuit, the worst-case dissipation rises to: PSYNC = [ISC(AVG)]2(1 + )RDS(ON)
1538 F12
LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
With the 0.033 sense resistor ISC(AVG) = 4A will result, increasing the Si4412DY dissipation to 950mW at a die temperature of 105C. CIN will require an RMS current rating of at least 1.5A at temperature and COUT will require an ESR of 0.03 for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR(IL) = 0.03(1.12A) = 34mVP-P PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1538-AUX/LTC1539. These items are also illustrated graphically in the layout diagram of Figure 13. Check the following in your layout: 1. Are the high current power ground current paths using or running through any part of signal ground? The LTC1438/LTC1438X/LTC1439 IC's have their sensitive pins on one side of the package. These pins include the signal ground for the reference, the oscillator input, the voltage and current sensing for both controllers and the low battery/comparator input. The signal ground area used on this side of the IC must return to the bottom plates of all of the output capacitors. The high current power loops formed by the input capacitors and the ground returns to the sources of the bottom N-channel MOSFETs, anodes of the Schottky diodes, and (-) plates of CIN, should be as short as possible and tied through a low resistance path to the bottom plates of the output capacitors for the ground return. 2. Do the LTC1538-AUX/LTC1539 SENSE- 1 and VOSENSE2 pins connect to the (+) plates of COUT? In adjustable applications, the resistive divider R1/R2 must be connected between the (+) plate of COUT and signal ground and the HF decoupling capacitor should be as close as possible to the LTC1538-AUX/LTC1539. 3. Are the SENSE- and SENSE+ leads routed together with minimum PC trace spacing? The filter capacitors between SENSE+ 1 (SENSE+ 2) and SENSE- 1 (SENSE- 2) should be as close as possible to the LTC1538-AUX/ LTC1539. 4. Do the (+) plates of CIN connect to the drains of the topside MOSFETs as closely as possible? This capacitor provides the AC current to the MOSFETs. 5. Is the INTVCC decoupling capacitor connected closely between INTVCC and the power ground pin? This capacitor carries the MOSFET driver peak currents. 6. Keep the switching nodes, SW1 (SW2), away from sensitive small-signal nodes. Ideally the switch nodes should be placed at the furthest point from the LTC1538-AUX/LTC1539. 7. Use a low impedance source such as a logic gate to drive the PLLIN pin and keep the lead as short as possible. PC BOARD LAYOUT SUGGESTIONS Switching power supply printed circuit layouts are certainly among the most difficult analog circuits to design. The following suggestions will help to get a reasonably close solution on the first try. The output circuits, including the external switching MOSFETs, inductor, secondary windings, sense resistor, input capacitors and output capacitors all have very large voltage and/or current levels associated with them. These components and the radiated fields (electrostatic and/or electromagnetic) must be kept away from the very sensitive control circuitry and loop compensation components required for a current mode switching regulator. The electrostatic or capacitive coupling problems can be reduced by increasing the distance from the radiator, typically a very large or very fast moving voltage signal. The signal points that cause problems generally include: the "switch" node, any secondary flyback winding voltage and any nodes which also move with these nodes. The switch, MOSFET gate, and boost nodes move between VIN and Pgnd each cycle with less than a 100ns transition time. The secondary flyback winding output has an AC signal component of - VIN times the turns ratio of the transformer, and also has a similar < 100ns transition time. The feedback control input signals need to have less than a few millivolts of noise in order for the regulator to perform properly. A rough calculation shows that 80dB of isolation at 2MHz is required from the switch node for low noise switcher operation. The situation is worse by a factor of the
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION
turns ratio for the secondary flyback winding. Keep these switch-node-related PC traces small and away from the "quiet" side of the IC (not just above and below each other on the opposite side of the board). The electromagnetic or current-loop induced feedback problems can be minimized by keeping the high ACcurrent (transmitter) paths and the feedback circuit (receiver) path small and/or short. Maxwell's equations are at work here, trying to disrupt our clean flow of current and voltage information from the output back to the controller input. It is crucial to understand and minimize the susceptibility of the control input stage as well as the more obvious reduction of radiation from the high-current output stage(s). An inductive transmitter depends upon the frequency, current amplitude and the size of the current loop to determine the radiation characteristic of the generated field. The current levels are set in the output stage once the input voltage, output voltage and inductor value(s) have been selected. The frequency is set by the outputstage transition times. The only parameter over which we have some control is the size of the antenna we create on the PC board, i.e., the loop. A loop is formed with the input capacitance, the top MOSFET, the Schottky diode, and the path from the Schottky diode's ground connection and the input capacitor's ground connection. A second path is formed when a secondary winding is used comprising the secondary output capacitor, the secondary winding and the rectifier diode or switching MOSFET (in the case of a synchronous approach). These "loops" should be kept as small and tightly packed as possible in order to minimize their "far field" radiation effects. The radiated field produced is picked up by the current comparator input filter circuit(s), as well as by the voltage feedback circuit(s). The current comparator's filter capacitor placed across the sense pins attenuates the radiated current signal. It is important to place this capacitor immediately adjacent to the IC sense pins. The voltage sensing input(s) minimizes the inductive pickup component by using an input capacitance filter to SGND. The capacitors in both case serve to integrate the induced current, reducing the susceptibility to both the "loop" radiated magnetic fields and the transformer or inductor leakage fields. The capacitor on INTVCC acts as a reservoir to supply the high transient currents to the bottom gates and to recharge the boost capacitor. This capacitor should be a 4.7F tantalum capacitor placed as close as possible to the INTVCC and PGND pins of the IC. Peak current driving the MOSFET gates exceeds 1A. The power ground pin of the IC, connected to this capacitor, should connect directly to the lower plates of the output capacitors to minimize the AC ripple on the INTVCC IC power supply. The previous instructions will yield a PC layout which has three separate ground regions returning separately to the bottom plates of the output capacitors: a signal ground, a MOSFET gate/INTVCC ground and the ground from the input capacitors, Schottky diode and synchronous MOSFET. In practice, this may produce a long power ground path from the input and output capacitors. A long, low resistance path between the input and output capacitor power grounds will not upset the operation of the switching controllers as long as the signal and power grounds from the IC pins does not "tap in" along this path.
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LTC1538-AUX/LTC1539
APPLICATIONS INFORMATION U W U U
RLP 10k CSS 0.1F
CLP 0.01F
1 1000pF CC1B 220pF CC1A 1000pF RC1 10k 1000pF INTVCC 2 3 4 5 COSC VIN 100k 6 7 8 9 CC2B 470pF RC2 10k CC2A 1000pF INT VCC 100pF 56pF 22pF OUTPUT DIVIDER REQUIRED WITH VPROG OPEN 1000pF 10 11 12 13 14 15 16 17 CSS 0.1F 18
RUN/SS1 SENSE
+1
PLL LPF PLLIN
36 35 34 33 32 31 30 29 28 27 26 25 24 23
EXT CLOCK CB1 0.1F
SENSE - 1 BOOST 1 VPROG1 ITH1 POR2 LTC1539 COSC SGND LBI LBO SFB1 ITH2 VPROG2 VOSENSE2 SENSE - 2 VIN BG1 INTVCC PGND BG2 EXTVCC TGS2 SW2 TGL2 TGL1 SW1 TGS1
M1
+
L1
CIN1
M3 RSENSE1 DB1 M2 D1
+
VOUT1 COUT1
+
VIN
+
GROUND PLANE
DB2 M6 L2 RSENSE2
M4
SENSE+ 2 BOOST 2 RUN/SS2 AUXDR AUXON AUXFB
21 20 19 CB2 0.1F
220pF 10
10
NOT ALL PINS CONNECTED FOR CLARITY BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 13. LTC1539 Physical Layout Diagram
+
22
CIN2
+
+
M5
D2
COUT2
VOUT2
-
1538 F13
+
4.7F
-
-
25
TYPICAL APPLICATIONS
2 RUN/SS1 SENSE + 1 VIN BG1 25 M1B 0.1F MBRS140T3 CMDSH-3 SENSE - 1 26 T1 1: 1.8 0.033 SW1 27 3 4
1N4148
100
VOUT1 5V/3A
LTC1538-AUX/LTC1539
1000pF
100
1000pF
+
100F 10V x2 GND
+
10k 6 23 M3 VOUT1 CMDSH-3 MBRS140T3 0.033 22 21 20 19 M2 18 17 0.1F 16 15
1000pF
5 VPROG1 INTVCC 24
4.7F 16V 100F 10V, x 2
220pF
56pF
VOUT2 3.3V/3.5A
22F 35V x2
470pF
1000pF
10k 22pF
10H SUMIDA CDRH125-100MC
392k, 1%
221k, 1%
1000pF
56pF
220pF
47k 2N2905
0.1F 10 10
ITH1 PGND LTC1538-AUX 7 COSC BG2 8 SGND EXTVCC 9 SFB1 SW2 10 ITH2 TGL2 11 VOSENSE2 BOOST 2 12 SENSE - 2 AUXON 13 SENSE + 2 AUXFB 14 RUN/SS2 AUXDR
R6 1M 1% VOUT3 12V
+
4.7F
R5 90.9k 1%
1k
1N4148 5V STANDBY
1538 TA01 VIN 5.2 TO 28V: SWITCHING FREQUENCY = 200kHz T1: DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID M1A, M1B = SILICONIX Si4936DY M2, M3 = SILICONIX Si4412DY ALL INPUT AND OUTPUT CAPACITORS ARE AVX-TPS SERIES
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+
26
LTC1538-AUX 5V/3A, 3.3V/3.5A, 12V/0.2A Regulator
10 0.1F 35V x2
+ 22F
MBRS1100T3
VIN 5.2V TO 28V
0.1F 1 BOOST1 TGL1 28 M1A
+
10F 25V
LTC1539 High Efficiency Low Noise 5V/20mA Standby, 5V/3A, 3.3V/3.5A and 12V/200mA Regulator
VIN 6V TO 28V
RLP 10k EXT CLOCK 100 1 RUN/SS1 SENSE + 1 PLLIN 0.1F 10F + 25V T1* 1: 1.8 VOUT1 5V/3A BOOST 1 TGL1 SW1 TGS1 LTC1539 COSC VIN BG1 INTVCC PGND BG2 EXTVCC TGS2 M6 SW2 TGL2 21 20 19 0.1F VOUT1 22 23 M4 24 25 L2 10H D4 CMDSH-3 26 M5 27 D3 MBRS140T3 COUT2 220F 10V 29 M2 4.7F 16V D1 MBRS140T3 SGND LBI LBO SFB1 ITH2 VPROG2 VOSENSE2 SENSE - 2 SENSE+ 2 BOOST 2 RUN/SS2 AUXON AUXFB AUXDR 30 D2 CMDSH-3 31 M3 RSENSE1 0.03 32 33 M1 34 SENSE - 1 VPROG1 ITH1 POR2 35 PLL LPF 36 2 1000pF 3 4 5 POR2 6 100k 7 INTVCC 100
CLP 0.01F
CSS1 0.1F
+
D7 MBRS1100T3
1000pF
CIN1 22F 35V x2
TYPICAL APPLICATIONS
CC1 1000pF
RC1 10k
COSC 56pF
+
8
100pF 10 100k 11 12 13 14 15 1000pF 16 17 CSS2 0.1F 10 18 CC2 1000pF
390k, 1% LBO
CC2A 470pF
RC 10k
1k
D6 1N4148
10
5V STANDBY
LTC1538-AUX/LTC1539
* T1 = DALE LPE6562-A262 OR BH ELECTRONICS #501-0657 M1, M2, M4, M5 = IRF7403 M3, M6 = IRLML2803 L2 = SUMIDA CDRH125-100MC ALL INPUT CAPACITORS ARE AVX-TPS SERIES ALL OUTPUT CAPACITORS ARE AVX-TPSV LEVEL II SERIES
+
220pF
+
+
110k, 1% 9 28
COUT1 220F 10V
VOUT2 3.3V 3.5A RSENSE2 0.03 CIN2 22F 35V x2
47k Q1 MMBT 2907
R6 1M 1% VOUT2 12V 200mA 4.7F 25V
+
R5 90.9k 1%
1538 TA02
U
CC1A 220pF
D5 1N4148
27
TYPICAL APPLICATIONS
3 SENSE - 1 BOOST 1 TGL1 SW1 TGS1 LTC1539 COSC VIN BG1 INTVCC PGND BG2 EXTVCC TGS2 M6 SW2 TGL2 21 20 19 0.1F VOUT1 22 23 M4 24 25 D4 CMDSH-3 26 M5 27 D3 MBRS140T3 COUT2 470F 10V 29 M2 SGND LBI LBO SFB1 ITH2 VPROG2 VOSENSE2 SENSE - 2 SENSE+ 2 BOOST 2 AUXON AUXFB RUN/SS2 AUXDR 30 D2 CMDSH-3 D1 MBRS140T3 31 10 M3 RSENSE1 0.025 32 33 M1 24V VPROG1 ITH1 POR2 34 4 5
LTC1538-AUX/LTC1539
T1* 9H 1:3.74 VOUT1 3.3V/4A
3.3F + 35V
CC1A, 220pF POR2 6 100k 7 8
CC1, 1000pF
RC1 10k
COSC 56pF
+
100pF LBO 10 100k 11 12 100pF 13 14 22pF 15 16 17 CSS2 0.1F 10 10 18 1000pF CC2 1000pF
390k, 1%
CC2A 470pF
RC 10k
56pF
L2 10H
RSENSE2 0.02
110k 1%
OUTPUT DIVIDER 121k REQUIRED WITH 1% VPROG OPEN DC
220pF
* T1 = DALE LPE-6562-A214 M1, M2, M4, M5 = Si4412DY M3, M6 = IRLML2803 L2 = SUMIDA CDRH127-100MC INPUT CAPACITORS ARE AVX-TPS SERIES OUTPUT CAPACITORS ARE AVX-TPSV LEVEL II SERIES
5V STANDBY
+
+
+
110k, 1% 9 28 0.1F 4.7F 16V
COUT1 330F 10V
VOUT2 2.5V 5A CIN2 22F 35V x2
47k MMBT2907 ALTI VOUT2 12V 200mA 4.7F 25V
R6 1M 1%
+
R5 90.9k 1%
1538 TA03
U
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VIN 4V TO 28V 100 100 EXT CLOCK
LTC1539 High Efficiency 5V/20mA Standby, 3.3V/2.5V Regulator with Low Noise 12V Linear Regulator
RLP 10k
CLP 0.01F
CSS1 0.1F
+
1 RUN/SS1 SENSE 0.1F
+1
1000pF 2 PLLIN 35 1000pF
PLL LPF
36 MBRS1100T3
CIN1 22F 35V x2
LTC1539 5-Output High Efficiency Low Noise 5V/3A, 3.3V/3A, 2.9V/2.6A, 12V/200mA, 5V/20mA Notebook Computer Power Supply (See PCB LAYOUT AND FILM for Layout of Schematic)
R18, 100 C13, 1000pF R19, 100 VIN (28V MAX) 1 RUN/SS1 SENSE + 1 PLLIN 34 M1A D6 CMDSH-3 R10 R5 4.7k 33 32 31 30 29 D2 CMDSH-3 M1B D1 MBRS140 C18, 0.01F M7 SENSE - 1 BOOST 1 VPROG1 TGL1 SW1 TGS1 LTC1539 COSC SGND BG1 INTVCC PGND BG2 EXTVCC TGS2 SW2 TGL2 21 20 VOUT1 5V STANDBY ( 20mA) C12 6.8nF R9 47k R1 27 Q1 2N2907 R2 100 R8, 316k,1% C17, 22pF Q1 = MOTOROLA, MMBT2907ALT1 Q2 = ZETEX, FZT849 T1 = DALE, LPE-6562-A236 L2 = SUMIDA, CDRH127-100MC ALL INPUT AND OUTPUT CAPACITORS ARE AVX-TPS SERIES R11 10 Q2 ZETEX FZT849 19 C27, 0.1F 5V/20mA STANDBY 22 23 M4 24 25 D4 CMDSH-3 L2 10H 26 M5 D3 MBRS140 27 C16, C19 100F 10V LBI LBO SFB1 ITH2 VPROG2 VOSENSE2 SENSE 2 SENSE+ 2 BOOST 2 RUN/SS2 AUXON AUXFB AUXDR
-
C15 1000pF
C14, 0.1F PLL LPF 35 R22 10 C20 0.1F 36 2 3 4 5 ITH1 POR2 VIN POR2 7 8 LB1 LBO 11 12 13 14 15 16 17 18 R21 10 R20 10 10 6 C28, C29 100F 10V T1* 10H 1:1.42
+
TYPICAL APPLICATIONS
R3 100k 1%
D5 MMBD914L
+
VOUT3 12V 120mA C4 3.3F 25V
C8 220pF
C2 1000pF
INTVCC
R13, 10k
C3 56pF
VOUT1 5V 3A
+
R15 10k
R4 11.3k 1%
R12 0.02 1W
C9 220pF
R12 1k
C10, 1000pF
C11 0.1F
D7 MMBD914L
R7 221k 1%
LTC1538-AUX/LTC1539
VIN 5.2V TO 28V: SWITCHING FREQUENCY = 200kHz 5V/3A, 3.3V/3A, 2.9V/1A, 2.6A PEAK, LINEAR 12V/200mA M1 = SILICONIX, Si4936DY M4, M5 = SILICONIX, Si4412DY M3, M6 = IRLML2803 M7 = INTERNATIONAL RECTIFIER, IRLL014
+
+
C7, 470pF
C6, 1000pF
+
9 28 C24, 4.7F, 16V
C23, 0.1F
C25, C26 22F 35V
VOUT2 3.3V 3A
+
C5 330F 6.3V
+
OPTIONAL 330F 6.3V
1538 TA04
VLDO 2.9V/1A 2.6A PEAK
U
C1, C21 C22 22F 35V
29
LTC1538-AUX/LTC1539
PCB LAYOUT A D FIL
Silkscreen Top
Copper Layer 1
Copper Layer 3
30
W
U
(Gerber files for this circuit board are available. Call the LTC factory.)
Silkscreen Bottom
Copper Layer 2 Ground Plane
Copper Layer 4
LTC1538-AUX/LTC1539
PACKAGE DESCRIPTION U
Dimensions in inches (millimeters) unless otherwise noted. G Package 28-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
0.397 - 0.407* (10.07 - 10.33) 28 27 26 25 24 23 22 21 20 19 18 17 16 15
0.301 - 0.311 (7.65 - 7.90)
1 2 3 4 5 6 7 8 9 10 11 12 13 14 0.205 - 0.212** (5.20 - 5.38) 0.068 - 0.078 (1.73 - 1.99)
0 - 8
0.005 - 0.009 (0.13 - 0.22)
0.022 - 0.037 (0.55 - 0.95)
0.0256 (0.65) BSC
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.010 - 0.015 (0.25 - 0.38)
0.002 - 0.008 (0.05 - 0.21)
G28 SSOP 0694
GW Package 36-Lead Plastic SSOP (Wide 0.300)
(LTC DWG # 05-08-1642)
0.602 - 0.612* (15.290 - 15.544) 36 35 34 33 32 31 30 29 28 27 26 25 24 23 22 21 20 19
0.400 - 0.410 (10.160 - 10.414)
0.292 - 0.299** (7.417 - 7.595) 0.010 - 0.016 x 45 (0.254 - 0.406) 0 - 8 TYP
1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 0.097 - 0.104 (2.463 - 2.641) 0.090 - 0.094 (2.286 - 2.387)
0.009 - 0.012 (0.231 - 0.305)
0.031 0.012 - 0.017 (0.800) (0.304 - 0.431) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.024 - 0.040 (0.610 - 1.016)
0.005 - 0.012 (0.127 - 0.305)
GW36 SSOP 0795
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC1538-AUX/LTC1539
TYPICAL APPLICATION
3.3V to 2.9V at 3A Low Noise Linear Regulator
5V 6.8nF 47k 27 3.3V Q1 MMBT2907ALTI 100 AUXDR LTC1539 AUXFB 2.9V ON/OFF AUXON 221k 1% 22pF 316k 1% 10 ZETEX FZT849 (SURFACE MOUNT) 2.9V 3A
RELATED PARTS
PART NUMBER LTC1142/LTC1142HV LTC1148/LTC1148HV LTC1159 LT 1375/LT1376 LTC1430 LTC1435
(R)
DESCRIPTION Dual High Efficiency Synchronous Step-Down Switching Regulators High Efficiency Step-Down Switching Regulator Controllers High Efficiency Step-Down Switching Regulator Controller 1.5A, 500kHz Step-Down Switching Regulators High Power Step-Down Switching Regulator Controller Single High Efficiency Low Noise Switching Regulator Controller
LTC1436/LTC1436-PLL/ High Efficiency Low Noise Synchronous Step-Down LTC1437 Switching Regulator Controllers LTC1438 LTC1439/LTC1438X Dual, Synchronous Controller with Power-On Reset and an Extra Comparator Dual Synchronous Controller with Power-On Reset, Extra Linear Controller, Adaptive Power, Synchronization, Auxiliary Regulator and an Extra Uncommited Comparators Constant-Voltage/Constant-Current Battery Charger
LT1510
32
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 q FAX: (408) 434-0507 q TELEX: 499-3977
U
+
330F x2
1538 TA05
COMMENTS Dual Synchronous, VIN 20V Synchronous, VIN 20V Synchronous, VIN 40V, For Logic Threshold FETs High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch High Efficiency 5V to 3.3V Conversion at Up to 15A 16-Pin Narrow SO and SSOP Packages Full-Featured Single Controllers Shutdown Current < 30A Shutdown Current < 30A, Power-On Reset Eliminated
1.3A, Li-Ion, NiCd, NiMH, Pb-Acid Charger
LT/GP 0896 7K * PRINTED IN USA
(c) LINEAR TECHNOLOGY CORPORATION 1996

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